Electronic current sensing and measurement is utilized in a wide variety of electronic devices. Current sensing and measurement methods and devices can be divided into two basic modes, voltage-based (indirect) and current-based (direct).
FIG. 1 is a simplified schematic of a typical voltage-based current sensing and measurement circuit 100. A current measurement resistor Rm 102 is placed in series with a load (not shown) in which the current is to be measured. A differential amplifier 104 is utilized to measure the voltage drop across Rm and the current is computed from the measured voltage drop and the known value of Rm.
This technique described above with respect to voltage-based current sensing and measurement circuit 100 has a number of drawbacks for measurement of large currents or current ranges having a large dynamic range (e.g., the range of the smallest current to be measured to the largest current to be measured). For large currents (e.g. on the order of several amperes) Rm needs to be as small as possible to minimize parasitic voltage drop and dissipated power. Manufacturing very low resistance values accurately is very difficult and expensive, however, particularly if it must be integrated on a monolithic integrated circuit. For very small currents, the limit of measurement will be determined by the D.C. parameters of differential amplifier 104, particularly the amplifier's offset voltage and, to a lesser extent, the input offset currents. As a result, the dynamic range will be limited to three or four orders of magnitude.
Current-based current sensing and measurement apparatus typically rely on what is commonly known as a “current” mirror. FIG. 2 is a simplified schematic 200 of a bipolar current mirror 200 that uses matched transistors to create an “image” current, or scaled replica, of the current to be measured, Iin. With bipolar current mirror 200, the current to be measured flows through “n” matched transistors 204, all having common emitter, base, and collector connections. Although three transistors 204 are illustrated in FIG. 2, n can be any number. Matching of the emitter base voltage characteristics assures that the current to be measured is equally shared by all the transistors 204. With the bipolar current mirror 200, an additional matched transistor 202 sharing the emitter and base connections of transistors 204 is employed to create the mirrored current Im, which is approximately 1/n of the current Iin.
One major drawback of bipolar current mirroring is excessive power dissipation at high current values. Since the typical emitter-base voltages for bipolar transistors are on the order of 0.6 to 0.7 volts, a current level of, for example, 1 ampere will result in a power dissipation of 600 to 700 mW. This high power dissipation creates difficulties for monolithic circuitry (e.g. integrated circuits), requiring expensive packaging, large dies sizes, and perhaps external heat sinking. As a result, current limits for bipolar current mirrors are typically no more than about 10 mA.
U.S. Pat. No. 6,888,401, incorporated herein by reference, describes a MOSFET-technology current mirror. FIG. 3 is a schematic of a MOSFET current mirror 300 utilizing matched MOSFETs (metal oxide semiconductor field effect transistors) 302 and 304 as described therein. The current Iin to be measured flows through MOSFET 302, which is designed to handle M times the current of MOSFET 304, at the same gate-to-source voltage. Thus, a current Iin flowing through MOSFET 302 induces a mirror current Im=Iin/M in MOSFET 304.
In FIG. 3, operational amplifier 308, in conjunction with bias control block 312, sets the output voltage of the MOSFET current mirror 300 by biasing the gate voltage of MOSFETs 302 and 304 such that the drain-to-source voltage of MOSFET 302 remains constant. This assures minimum power dissipation while keeping MOSFET 302 in linear operation. Operational amplifier (“op-amp”) 310 and MOSFET 306 keep the drain-to-source voltages of MOSFETs 302 and 304 equal to each other, within the error of the input offset voltage of op-amp 310. This assures tight matching of MOSFETs and reduces errors that may result if the drain-to-source voltages are allowed to vary.
While the performance of the circuit of FIG. 3 is an improvement over the voltage-based version of FIG. 1 and the bipolar current mirror of FIG. 2, it still exhibits a degree of inaccuracy at large M values of about 1000 or greater. For these large M values, accuracy is typically limited to 8-10%, primarily due to monolithic circuit layout issues which affect the matching of MOSFETs 302 and 304, proximity effects, interconnect resistance, and package stress.
Improvements in the accuracy of the MOSFET current mirror shown in FIG. 3 can be made if the gain M can be reduced. At reduced gains, the matching of MOSFETs 302 and 304 can be more precise, and the current measurement accuracy can be significantly improved. However, reducing M can create other problems, particularly when trying to measure currents on the order of a few amperes.
These and other limitations of the prior art will become apparent to those of skill in the art upon a reading of the following descriptions and a study of the several figures of the drawing.